1. Field of the Invention
This invention generally relates to capacitive measurement circuits and methods and particularly to circuits and methods for measuring small values of capacitance with good rejection of circuit and ambient noise.
2. Description of Related Art
Capacitive sensors have many uses. These include measurements of pressure, humidity, motion, rotation, material composition, and proximity variables. In practice, a variable to be sensed is converted to a variable capacitance, this variable capacitance is measured, and its value is observed directly or processed by computer. The capacitance levels may range from a small fraction of a picofarad to many picofarads.
A typical problem requiring measurement of small capacitance in the presence of noise is to detect the proximity of human hand, for example a hand about to be trapped in a closing automobile window or caught in a machine. Systems are available for this purpose that excite a metal plate, perhaps 1″×10″, with an AC voltage of several volts and 1 kHz–1 MHz frequency, and measure the plate's capacitance to ground. As a hand nears, this capacitance increases by a very small value, typically less than one pF. As the environment is often electrically noisy, with nearby fluorescent lamps or radio transmitters, a critical specification is the circuit's noise rejection.
The extraordinary sensitivity of the technology is explored by Jones and Richard's seminal paper [Jones, R. V. and J. C. S. Richards, 1973, The Design and Some Applications of Capacitive Micrometers, Journal of Physics E: Scientific Instruments 6: 589–600]; this paper demonstrates a signal to noise ratio (SNR) of well over a million to one. Yet many capacitive sensor tasks are at the limit established by amplifier noise and would benefit by lower noise. Normal capacitive motion sensors have a resolution about the same as laser interferometers, considered the state of the precision motion measurement art; an order of magnitude improvement would allow even more precise metrology.
Two different circuits are commonly used to demodulate capacitive sensors for lowest noise:
1. Classic designs [Jones and Richards; Baxter, L. K., 1997, Capacitive Sensors. N.J.: IEEE Press, p.54] drive the capacitor to be measured with, say, a 5V square wave at 100 kHz, and measure the resulting current with a linear amplifier followed by a synchronous demodulator.
2. Improved charge-balance designs [Baxter, p. 55] drive the measured capacitor and a reference capacitor with opposite phases, with the drive voltage adjusted so that the resulting current is zero. A linear amplifier and synchronous demodulator are used here also.
Capacitive sensors that operate in a noisy environment are described in the following references:                U.S. Pat. No. 5,436,613 (1995) Ghosh et al.        U.S. Pat. No. 5,525,843 (1996) Howing        U.S. Pat. No. 5,722,686 (1998) Blackburn et al.        U.S. Pat. No. 5,744,968 (1998) Czarnocki et al.        U.S. Pat. No. 5,802,479 (1998) Kithil et al.        U.S. Pat. No. 6,158,768 (2000) Steffen, Jr. et al.        
Two-dimensional finger position sensors or touch panels for computer input often use capacitive sensing. U.S. Pat. No. 4,698,461 (1987) to Meadows et al. shows a capacitively-sensed touch panel that changes the circuit's operation frequency to avoid interfering noise sources; this added circuit complexity would not be needed if the circuit was intrinsically less susceptible to noise.
As described in Baxter, Larry K., Capacitive Sensors, supra, three well-known circuits for detecting a small capacitance are (1) the RC oscillator circuit, (2) the synchronous demodulator circuit, and (3) the charge transfer circuit. These circuits have different strengths and weaknesses, and an understanding of their operation is important for the understanding of the present invention.
FIG. 1 depicts a prior-art RC oscillator, such as shown in U.S. Pat. No. 6,307,385 (2001) to Tardif et al. This is a simple circuit in which frequency is proportional to the reciprocal of capacitance, as given by:f=K/RCx  (1)where K is a constant determined by the threshold voltage of a Schmitt trigger 2,
f is output frequency
R is resistance, here 50K ohms, and
Cx is the measured capacitance of variable capacitor 1.
The RC oscillator detects capacitance as a frequency variation. But it is very susceptible to noise. Interfering noise can be considered as either AC noise or impulse noise. AC noise typically is confined to a narrow frequency band, illustrated by AM radio transmitters or power line radiation. Impulse noise typically is confined to a narrow time slice, like switch noise, motor brush noise, or semiconductor lamp dimmer transients.
The RC oscillator of FIG. 1 is susceptible to both noise sources. First, AC noise coupling to the variable capacitance is directly added to the measurement output. This is a serious drawback, as most industrial sites have considerable noise at power frequencies and their harmonics, peaking at 50 or 60 Hz and decreasing towards 100 kHz. Also, impulse noise acts to increase the frequency by triggering the oscillator prematurely. This behavior is typical of any sense circuit that includes a comparator: an impulse just before the RC voltage reaches the comparator threshold triggers the cycle early, but an impulse just after the threshold is ignored. This imparts a DC offset that is not removed by a following low pass filter. Other circuits, like those depicted in FIGS. 2 and 4, do not have this behavior.
FIG. 2 discloses a charge transfer circuit of the type described in U.S. Pat. No. 4,345,167 (1982) to Calvin. It has low power dissipation and better noise rejection than the RC oscillator circuit in FIG. 1. In operation, semiconductor switch 5 normally connects capacitor 4, Cx, in parallel with a small stray capacitance 41, to the DC supply voltage 3, say 5V. Cx, charged to 5V, holds a charge Q=CV or 5 pC for a 1 pF Cx. The switch 5 is then momentarily connected to capacitor 6, Cs, for a uS or less, as shown in the timing diagram of FIG. 3. This transfers most of Cx's charge to Cs when SAMPLE is high. SAMPLE is advantageously set to the minimum time t0 that will allow full charge transfer.
As Cs is usually many times larger than Cx, say 100 times larger, the voltage on Cs increases by about 50 mV with each SAMPLE pulse. After switch 5 is cycled perhaps 20 times, the voltage on capacitor 6 is nearly 1V, and this voltage can be easily measured.
The output voltage Vomax is then read externally, coincident with the READ pulse of FIG. 3, then the reset switch 7 is momentarily connected to discharge Cs and the measurement cycle is repeated.
The timing diagram of FIG. 3 shows operation with just four charge transfer pulses. As the number of charge transfer pulses per read-reset operation increases, the noise rejection increases but the response time decreases.
This circuit has an important advantage of sampling speed. It is sensitive to noise only during the very short time interval when switch 5 is connected to capacitor 6, perhaps 20 nS for a fast switch. If t1 is chosen as 10 uS so the excitation frequency is 100 kHz, the circuit is open to noise only 0.2% of the time, and the noise rejection is 500×.
The short sample time possible with a low-cost CMOS switch contributes the noise rejection of a very fast excitation frequency without power-hungry amplifiers, and while using a low excitation frequency with its advantages in power and electromagnetic interference. This fast sample rate at low power, the inherent noise-reducing averaging across many samples, and the voltage gain without amplifiers make this an attractive circuit.
However, the charge transfer circuit does not reject AC noise very well. The narrow sampling window improves impulse noise performance considerably compared to the RC oscillator of FIG. 1, but the circuit is influenced by AC noise over a wide bandwidth. Noise frequencies of 60 Hz, for example, couple to Cx and appear directly in the output.
Another drawback of the simple circuit of FIG. 2 is that it is nonlinear, with an exponential transfer function
                              Vo          i                =                  Vs          (                      1            -                          ⅇ                                                                    -                    Cx                                    Cs                                ⁢                i                                              )                                    (        2        )            where i is the number of sample pulses.
Also in this circuit stray capacitance 41 from the sensed node of Cx to ground adds to the measured capacitance and hurts accuracy.
Many other charge transfer circuits are described in the literature, such as:                U.S. Pat. No. 5,451,940 (1995) Schneider et al.        U.S. Pat. No. 5,751,154 (1998) Tsugai        U.S. Pat. No 6,377,056 (2002) Hanzawa et al.However, none of these references uses AC excitation so each is susceptible to errors due to noise.        
The prior art synchronous demodulator circuit of FIG. 4 shows considerable improvement over the circuit of FIG. 1. The sensed capacitor 11 is excited with, for example, a square wave generator 8 at 100 kHz. This excitation signal can be produced by a logic gate. This 100 kHz signal also controls switch 15.
A reference capacitor 10 works with measured capacitance 11 as a voltage divider. A unity-gain, low-bias-current operational amplifier 13 buffers the very high capacitive impedance. This amplifier is preferably a FET-input type with a frequency response greater than 10 MHz, such as Analog Device's AD823. Some method of setting the DC level at the amplifier input is needed, such as the high-value resistor 12 or a momentary switch to ground (not shown).
Stray capacitance to ground, as with capacitor 41 of FIG. 2, can add to capacitor 11 and hurt the measurement accuracy. A prior-art solution is shown, where the sense node is shielded and the shield 9 is connected to the output of the unity gain amplifier. The stray node capacitance is converted from capacitance to ground to capacitance to the shield. Stray capacitance is then driven on both sides by the same voltage, no current can flow in it and it disappears from the circuit equation, except by adding to noise. This guard technique can be applied to any of the circuits of this patent.
The variable amplitude square wave at the output of amplifier 13 feeds the synchronous demodulator 14–15, where SPDT switch 15 is a high-speed CMOS switch available from many semiconductor manufacturers, such as the Maxim MAX4053 CMOS switch available from Maxim Integrated Products, Inc. of Sunnyvale, Calif. If the circuit and the switch are integrated on silicon, the switch can have improved performance and lower capacitance. The synchronous demodulator inverts alternate half cycles of the 100 kHz square wave, and the 100 kHz component of the resulting rectified signal is removed by low pass filter 16. FIG. 5 shows the excitation waveform and the variable-amplitude signal at the output of V13, and the input of the low pass filter 16, VLPI.
The filtered output measures the capacitance with the nonlinear equation:Vo=VsCx/(Cx+Cr   (3)
This circuit rejects impulse noise better than the RC oscillator but not as well as the charge transfer circuit of FIG. 2. It rejects AC noise better than either; it is sensitive to AC noise only if its frequency is near the 100 kHz carrier, specifically within a frequency band equal to twice the low pass filter's cutoff frequency. centered on 100 kHz. As the LPF bandwidth can be much smaller than 100 kHz, say 1 Hz, the synchronous demodulator can have a very narrow-band response that rejects AC noise. To see how this works, imagine a 60 Hz signal coupled to Cx. It appears at the input of the low pass filter 16 as a alternate-cycle modulation at a 100 kHz frequency, but the low pass filter will almost completely remove this high-frequency modulation and hence the 60 Hz component.
The low pass filter type can be selected to optimize noise rejection, with a simple RC low pass for AC noise or a median filter for impulse noise. For best noise rejection the excitation frequency should be very high, say 10 MHz, and the operational amplifier should have ten times this bandwidth for good stability. As the sample time is reduced and the number of sample pulses increases, noise rejection improves directly. A limitation of the synchronous demodulator circuit for low-noise applications is that this high frequency operation requires expensive, power-hungry components and may cause excessive electromagnetic radiation.